View AD8013ARZ-14_7557353.PDF datasheet online --- IC-ON-LINE

Datasheet File OCR Text:

rev. a a information furnished by analog devices is believed to be accurate and reliable. however, no responsibility is assumed by analog devices for its use, nor for any infringements of patents or other rights of third parties which may result from its use. no license is granted by implication or otherwise under any patent or patent rights of analog devices. single supply, low power, triple video amplifier features three video amplifiers in one package drives large capacitive load excellent video specifications (r l = 150 v ) gain flatness 0.1 db to 60 mhz 0.02% differential gain error 0.06 differential phase error low power operates on single +5 v to +13 v power supplies 4 ma/amplifier max power supply current high speed 140 mhz unity gain bandwidth (3 db) fast settling time of 18 ns (0.1%) 1000 v/ m s slew rate high speed disable function per channel turn-off time 30 ns easy to use 95 ma short circuit current output swing to within 1 v of rails applications lcd displays video line driver broadcast and professional video computer video plug-in boards consumer video rgb amplifier in component systems ad8013 pin configuration 14-pin dip & soic package 1 2 3 4 5 6 7 14 13 12 11 10 9 8 ad8013 out 2 ?n 2 +in 2 ? s +in 3 ?n 3 out 3 disable 1 disable 2 disable 3 +v s +in 1 ?n 1 out 1 product description the ad8013 is a low power, single supply, triple video amplifier. each of the three amplifiers has 30 ma of output current, and is optimized for driving one back terminated video load (150 w ) each. each amplifier is a current feedback amp- lifier and features gain flatness of 0.1 db to 60 mhz while offering frequency ?hz ?.5 1m 1g 10m normalized gain ?db 100m 0.2 0.1 0 ?.1 ?.2 ?.3 ?.4 g = +2 r l = 150 w v s = 5v v s = +5v fine-scale gain flatness vs. frequency, g = +2, r l = 150 w ? analog devices, inc., 1995 one technology way, p.o. box 9106, norwood, ma 02062-9106, u.s.a. tel: 617/329-4700 fax: 617/326-8703 differential gain and phase error of 0.02% and 0.06 . this makes the ad8013 ideal for broadcast and professional video electronics. the ad8013 offers low power of 4 ma per amplifier max and runs on a single +5 v to +13 v power supply. the outputs of each amplifier swing to within one volt of either supply rail to easily accommodate video signals. the ad8013 is unique among current feedback op amps by virtue of its large capacitive load drive. each op amp is capable of driving large capacitive loads while still achieving rapid settling time. for instance it can settle in 18 ns driving a resistive load, and achieves 40 ns (0.1%) settling while driving 200 pf. the outstanding bandwidth of 140 mhz along with 1000 v/ m s of slew rate make the ad8013 useful in many general purpose high speed applications where a single +5 v or dual power supplies up to 6.5 v are required. furthermore the ad8013s high speed disable function can be used to power down the amplifier or to put the output in a high impedance state. this can then be used in video multiplexing applications. the ad8013 is available in the industrial temperature range of C40 c to +85 c. 1 0 0% 100 9 0 500ns 500mv 5v channel switching characteristics for a 3:1 mux
ad8013Cspecifications model ad8013a conditions v s min typ max units dynamic performance bandwidth (3 db) no peaking, g = +2 +5 v 100 125 mhz no peaking, g = +2 5 v 110 140 mhz bandwidth (0.1 db) no peaking, g = +2 +5 v 50 mhz no peaking, g = +2 5 v 60 mhz slew rate 2 v step +5 v 400 v/ m s 6 v step 5 v 600 1000 v/ m s settling time to 0.1% 0 v to +2 v 5 v 18 ns 4.5 v step, c load = 200 pf 6 v 40 ns r load > 1 k w , r fb = 4 k w noise/harmonic performance total harmonic distortion f c = 5 mhz, r l = 1 k 5 v C76 dbc f c = 5 mhz, r l = 150 w 5 v C66 dbc input voltage noise f = 10 khz +5 v, 5 v 3.5 nv/ ? hz input current noise f = 10 khz (Ci in ) +5 v, 5 v 12 pa/ ? hz differential gain (r l = 150 w ) f = 3.58 mhz, g = +2 +5 v 1 0.05 % 5 v 0.02 0.05 % differential phase (r l = 150 w ) f = 3.58 mhz, g = +2 +5 v 1 0.06 degrees 5 v 0.06 0.12 degrees dc performance input offset voltage t min to t max +5 v, 5 v 2 5 mv offset drift 7 m v/ c input bias current (C) +5 v, 5 v 2 10 m a input bias current (+) t min to t max +5 v, 5 v 3 15 m a open-loop transresistance +5 v 650 800 k w t min to t max 550 k w 5 v 800 k 1.1 m w t min to t max 650 k w input characteristics input resistance +input 5 v 200 k w Cinput 5 v 150 w input capacitance 5 v 2 pf input common-mode voltage range 5 v 3.8 v +5 v 1.2 3.8 +v common-mode rejection ratio input offset voltage +5 v 52 56 db input offset voltage 5 v 52 56 db Cinput current +5 v, 5 v 0.2 0.4 m a/v +input current +5 v, 5 v 5 7 m a/v output characteristics output voltage swing r l = 1 k w v ol Cv ee 0.8 1.0 v v cc Cv oh 0.8 1.0 v r l = 150 w v ol Cv ee 1.1 1.3 v v cc Cv oh 1.1 1.3 v output current +5 v 30 ma 5 v 25 30 ma short-circuit current 5 v 95 ma capacitive load drive 5 v 1000 pf matching characteristics dynamic crosstalk g = +2, f = 5 mhz +5 v, 5 v 70 db gain flatness match f = 20 mhz 5 v 0.1 db dc input offset voltage +5 v, 5 v 0.3 mv Cinput bias current +5 v, 5 v 1.0 m a (@ t a = +25 8 c, r load = 150 v , unless otherwise noted) C2C rev. a
ad8013 model ad8013a conditions v s min typ max units power supply operating range single supply +4.2 +13 v dual supply 2.1 6.5 v quiescent current/amplifier +5 v 3.0 3.5 ma 5 v 3.4 4.0 ma 6.5 v 3.5 ma quiescent current/amplifier power down +5 v 0.25 0.35 ma 5 v 0.3 0.4 ma power supply rejection ratio input offset voltage v s = 2.5 v to 5 v 70 76 db Cinput current +5 v, 5 v 0.03 0.2 m a/v +input current +5 v, 5 v 0.07 1.0 m a/v disable characteristics off isolation f = 6 mhz +5 v, 5 v C70 db off output impedance g = +1 +5 v, 5 v 12 pf turn-on time 50 ns turn-off time 30 ns switching threshold Cv s + xv 1.3 1.6 1.9 v notes 1 the test circuit for differential gain and phase measurements on a +5 v supply is ac coupled. specifications subject to change without notice. C3C rev. a absolute maximum ratings 1 supply voltage . . . . . . . . . . . . . . . . . . . . . . . . . . 13.2 v total internal power dissipation 2 plastic (n) . . . . . . . . . 1.6 watts (observe derating curves) small outline (r) . . . . 1.0 watts (observe derating curves) input voltage (common mode) . . lower of v s or 12.25 v differential input voltage . . . . . . . . output 6 v (clamped) output voltage limit maximum . . . . . . . . . lower of (+12 v from Cv s ) or (+v s ) minimum . . . . . . . . . higher of (C12.5 v from +v s ) or (Cv s ) output short circuit duration . . . . . . . . . . . . . . . . . . . . observe power derating curves storage temperature range n and r package . . . . . . . . . . . . . . . . . . . C65 c to +125 c operating temperature range ad8013a . . . . . . . . . . . . . . . . . . . . . . . . . . C40 c to +85 c lead temperature range (soldering 10 sec) . . . . . . . . +300 c notes 1 stresses above those listed under absolute maximum ratings may cause permanent damage to the device. this is a stress rating only and functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. exposure to absolute maximum rating conditions for extended periods may affect device reliability. 2 specification is for device in free air: 14-pin plastic dip package: q ja = 75 c/watt 14-pin soic package: q ja = 120 c/watt ordering guide temperature package package model range description options ad8013an C40 c to +85 c 14-pin plastic dip n-14 ad8013ar-14 C40 c to +85 c 14-pin plastic soic r-14 ad8013ar-14-reel C40 c to +85 c 14-pin plastic soic r-14 ad8013ar-14-reel7 C40 c to +85 c 14-pin plastic soic r-14 ad8013achips C40 c to +85 c die form maximum power dissipation the maximum power that can be safely dissipated by the ad8013 is limited by the associated rise in junction temperature. the maximum safe junction temperature for the plastic encapsulated parts is determined by the glass transition temperature of the plastic, about 150 c. exceeding this limit temporarily may cause a shift in parametric performance due to a change in the stresses exerted on the die by the package. exceeding a junction temperature of 175 c for an extended period can result in device failure. while the ad8013 is internally short circuit protected, this may not be enough to guarantee that the maximum junction temper- ature is not exceeded under all conditions. to ensure proper operation, it is important to observe the derating curves. it must also be noted that in (noninverting) gain configurations (with low values of gain resistor), a high level of input overdrive can result in a large input error current, which may result in a significant power dissipation in the input stage. this power must be included when computing the junction temperature rise due to total internal power. maximum power dissipation ?watts ambient temperature ? c 2.5 2.0 0.5 ?0 90 ?0 ?0 ?0 0 10 20 30 40 50 60 70 80 1.5 1.0 ?0 t j = +150 c 14-pin dip package 14-pin soic maximum power dissipation vs. ambient temperature
ad8013 rev. a C4C metalization photo contact factory for latest dimensions. dimensions shown in inches and (mm). +in1 5 +v s 4 disable 3 3 2 disable 2 1 disable 1 14 out 2 ?n1 6 out1 7 out3 8 ?n3 9 10 +in3 11 ? s 12 +in2 13 ?n2 0.071 (1.81) 0.044 (1.13) caution esd (electrostatic discharge) sensitive device. electrostatic charges as high as 4000 v readily accumulate on the human body and test equipment and can discharge without detection. although the ad8013 features proprietary esd protection circuitry, permanent damage may occur on devices subjected to high energy electrostatic discharges. therefore, proper esd precautions are recom- mended to avoid performance degradation or loss of functionality. warning! esd sensitive device supply voltage ? volts 6 0 17 2 common-mode voltage range ? volts 3456 5 4 3 2 1 figure 1. input common-mode voltage range vs. supply voltage supply voltage ? volts 12 0 17 2 output voltage swing ?v p-p 3456 10 8 6 4 2 no load r l = 150 w figure 2. output voltage swing vs. supply voltage
C5C rev. a ad8013 load resistance ? w 10 8 0 10 10k 100 output voltage swing ?v p-p 1k 6 4 2 v s = 5v v s = +5v figure 3. output voltage swing vs. load resistance junction temperature ? c 12 9 6 ?0 140 ?0 supply current ?ma ?0 0 20 40 60 80 100 120 11 10 8 7 v s = 5v v s = +5v figure 4. total supply current vs. junction temperature supply voltage ? volts 11 7 supply current ?ma 9 8 10 17 23456 t a = +25 c figure 5. supply current vs. supply voltage junction temperature ? c 3 0 ? ?0 140 ?0 input bias current ?? ?0 0 20 40 60 80 100 120 2 1 ? ? ? b +i b figure 6. input bias current vs. junction temperature junction temperature ? c 2 ? ? ?0 140 ?0 input offset voltage ?mv ?0 0 20 40 60 80 100 120 1 0 ? ? v s = +5v v s = 5v figure 7. input offset voltage vs. junction temperature junction temperature ? c 140 130 80 ?0 140 ?0 short circuit current ?ma ?0 0 20 40 60 80 100 120 120 100 90 source sink v s = 5v figure 8. short circuit current vs. junction temperature
ad8013 rev. a C6C frequency ?hz 10 100k 1g 1m common-mode rejection ?db 10m 100m 70 60 20 50 40 30 v cm r r r r figure 12. common-mode rejection vs. frequency frequency ?hz 80 0 100k 1g 1m 10m 100m 70 power supply rejection ?db 60 10 +psr 20 30 40 50 ?sr v s = 5v figure 13. power supply rejection ratio vs. frequency frequency ?hz 120 40 100k 1g 1m transimpedance ?db 10m 100m 100 80 60 0 ?5 ?0 ?35 ?80 phase ?degrees 140 10k v s = 5v r l = 1k figure 14. open-loop transimpedance vs. frequency (relative to 1 w ) frequency ?hz 1k 100 0.01 100k 1g 1m closed-loop output resistance ? w 10m 100m 10 1 0.1 v s = 5v g = +2 figure 9. closed-loop output resistance vs. frequency frequency ?hz 100k 10k 10 1m 1g 10m output resistance ? w 100m 1k 100 figure 10. output resistance vs. frequency, disabled state frequency ?hz 1k 100 1 100 1m 1k voltage noise nv/ ? hz 10k 100k 10 1k 100 1 10 current noise pa/ ? hz noninverting i inverting i v noise figure 11. input current and voltage noise vs. frequency
C7C rev. a ad8013 frequency ?hz 1k 100m 10k harmonic distortion ?dbc 100k 1m 10m ?0 ?0 ?0 ?0 ?0 ?0 ?0 ?00 ?10 ?20 g = +2 v o = 2v p-p v s = 5v 2nd r l = 150 w 2nd r l = 1k w 3rd r l = 1k w 3rd r l = 150 w figure 15. harmonic distortion vs. frequency output step size ?v p-p 18 234567 1800 1600 slew rate ? v/? 800 600 400 200 1200 1000 1400 v s = 5v r l = 500 w g = +10 g = ? g = +2 g = +1 figure 16. slew rate vs. output step size 10 0% 100 90 20ns 2v 2v v in v out figure 17. large signal pulse response, gain = +1, (r f = 2 k w , r l = 150 w , v s = 5 v) frequency ?hz 1m 1g 10m closed-loop gain (normalized) ?db 100m ? +1 0 ? ? ? ? ? 0 ?0 ?80 ?70 phase shift ?degrees g = +1 r l = 150 w v s = 5v v s = +5v v s = +5v v s = 5v gain phase figure 18. closed-loop gain and phase vs. frequency, g = +1, r l = 150 w supply voltage ? volts 2000 1.5 7.5 2.5 slew rate ? v/? 3.5 4.5 5.5 6.5 1800 1200 600 400 200 1600 1400 1000 800 g = +10 g = ? g = +2 g = +1 figure 19. maximum slew rate vs. supply voltage 10 0% 100 90 20ns 500mv 500mv v in v out figure 20. small signal pulse response, gain = +1, (r f = 2 k w , r l = 150 w , v s = 5 v)
ad8013 rev. a C8C 10 0% 100 90 20ns 50mv 500mv v in v out figure 21. large signal pulse response, gain = +10, r f = 301 w , r l = 150 w , v s = 5 v) frequency ?hz 1m 1g 10m closed-loop gain (normalized) ?db 100m ? +1 0 ? ? ? ? ? 0 ?0 ?80 ?70 phase shift ?degrees g = +10 r l = 150 w v s = 5v v s = +5v v s = +5v v s = 5v gain phase figure 22. closed-loop gain and phase vs. frequency, g = +10, r l = 150 w 10 0% 100 90 20ns 50mv 500mv v in v out figure 23. small signal pulse response, gain = +10, (r f = 301 w , r l = 150 w , v s = 5 v) 10 0% 100 90 20ns 2v 2v v in v out figure 24. large signal pulse response, gain = C1, (r f = 698 w , r l = 150 w , v s = 5 v) frequency ?hz 1m 1g 10m closed-loop gain (normalized) ?db 100m ? +1 0 ? ? ? ? ? 0 90 180 ?0 phase shift ?degrees g = ? r l = 150 w v s = 5v v s = +5v v s = +5v v s = 5v gain phase figure 25. closed-loop gain and phase vs. frequency, g = C1, r l = 150 w 10 0% 100 90 20ns 500mv 500mv v in v out figure 26. small signal pulse response, gain = C1, (r f = 698 w , r l = 150 w , v s = 5 v)
C9C rev. a ad8013 frequency ?hz 1m 1g 10m closed-loop gain (normalized) ?db 100m ? +1 0 ? ? ? ? ? 180 90 0 ?0 phase shift ?degrees g = ?0 r l = 150 w v s = 5v v s = +5v v s = +5v v s = 5v gain phase figure 27. closed-loop gain and phase vs. frequency, g = C10, r l = 150 w to estimate the C3 db bandwidth for closed-loop gains of 2 or greater, for feedback resistors not listed in the following table, the following single pole model for the ad8013 may be used: acl . g 1 + sc t ( r f + gn rin ) where: c t = transcapacitance > 1 pf r f = feedback resistor g = ideal closed loop gain gn = 1 + r f r g ? ? ? ? = noise gain rin = inverting input resistance > 150 w acl = closed loop gain the C3 db bandwidth is determined from this model as: f 3 . 1 2 p c t ( r f + gn rin ) this model will predict C3 db bandwidth to within about 10% to 15% of the correct value when the load is 150 w and v s = 5 v. for lower supply voltages there will be a slight decrease in bandwidth. the model is not accurate enough to predict either the phase behavior or the frequency response peaking of the ad8013. it should be noted that the bandwidth is affected by attenuation due to the finite input resistance. also, the open-loop output resistance of about 12 w reduces the bandwidth somewhat when driving load resistors less than about 250 w . (bandwidths will be about 10% greater for load resistances above a few hundred ohms.) table i. C3 db bandwidth vs. closed-loop gain and feedback resistor, r l = 150 w (soic) v s C volts gain r f C ohms bw C mhz 5 +1 2000 230 +2 845 (931) 150 (135) +10 301 80 C1 698 (825) 140 (130) C10 499 85 +5 +1 2000 180 +2 887 (931) 120 (130) +10 301 75 C1 698 (825) 130 (120) C10 499 80 driving capacitive loads when used in combination with the appropriate feedback resistor, the ad8013 will drive any load capacitance without oscillation. the general rule for current feedback amplifiers is that the higher the load capacitance, the higher the feedback resistor required for stable operation. due to the high open-loop transresistance and low inverting input current of the ad8013, the use of a large feedback resistor does not result in large closed- loop gain errors. additionally, its high output short circuit current makes possible rapid voltage slewing on large load capacitors. for the best combination of wide bandwidth and clean pulse response, a small output series resistor is also recommended. table ii contains values of feedback and series resistors which result in the best pulse responses. figure 29 shows the ad8013 driving a 300 pf capacitor through a large voltage step with virtually no overshoot. (in this case, the large and small signal pulse responses are quite similar in appearance.) general the ad8013 is a wide bandwidth, triple video amplifier that offers a high level of performance on less than 4.0 ma per amplifier of quiescent supply current. the ad8013 uses a proprietary enhancement of a conventional current feedback architecture, and achieves bandwidth in excess of 200 mhz with low differential gain and phase errors, making it an extremely efficient video amplifier. the ad8013s wide phase margin coupled with a high output short circuit current make it an excellent choice when driving any capacitive load. high open-loop gain and low inverting input bias current enable it to be used with large values of feedback resistor with very low closed-loop gain errors. it is designed to offer outstanding functionality and performance at closed-loop inverting or noninverting gains of one or greater. choice of feedback & gain resistors because it is a current feedback amplifier, the closed-loop band- width of the ad8013 may be customized using different values of the feedback resistor. table i shows typical bandwidths at different supply voltages for some useful closed-loop gains when driving a load of 150 w . the choice of feedback resistor is not critical unless it is important to maintain the widest, flattest frequency response. the resistors recommended in the table are those (chip resistors) that will result in the widest 0.1 db bandwidth without peaking. in applications requiring the best control of bandwidth, 1% resistors are adequate. package parasitics vary between the 14-pin plastic dip and the 14-pin plastic soic, and may result in a slight difference in the value of the feedback resistor used to achieve the optimum dynamic performance. resistor values and widest bandwidth figures are shown in parenthesis for the soic where they differ from those of the dip. wider bandwidths than those in the table can be attained by reducing the magnitude of the feedback resistor (at the expense of increased peaking), while peaking can be reduced by increasing the magnitude of the feedback resistor. increasing the feedback resistor is especially useful when driving large capacitive loads as it will increase the phase margin of the closed-loop circuit. (refer to the section on driving capacitive loads for more information.)
ad8013 rev. a C10C 4 +v s ad8013 1.0? 0.1? 11 1.0? 0.1? ? s r g r t v in 15 w c l v o r f r s figure 28. circuit for driving a capacitive load table ii. recommended feedback and series resistors vs. capacitive load and gain r s C ohms c l C pf r f C ohms g = 2 g 3 3 20 2k 25 15 50 2k 25 15 100 3k 20 15 200 4k 15 15 300 6k 15 15 3 500 7k 15 15 10 0% 100 90 50ns 500mv 1v v in v out figure 29. pulse response driving a large load capacitor. c l = 300 pf, g = +2, r f = 6k, r s = 15 w overload recovery the three important overload conditions are: input common- mode voltage overdrive, output voltage overdrive, and input current overdrive. when configured for a low closed-loop gain, the amplifier will quickly recover from an input common- mode voltage overdrive; typically in under 25 ns. when con- figured for a higher gain, and overloaded at the output, the recovery time will also be short. for example, in a gain of +10, with 15% overdrive, the recovery time of the ad8013 is about 20 ns (see figure 30). for higher overdrive, the response is somewhat slower. for 6 db overdrive, (in a gain of +10), the recovery time is about 65 ns. 10 0% 100 90 50ns 500mv 5v v in v out figure 30. 15% overload recovery, g = +10 (r f = 300 w , r l = 1 k w , v s = 5 v) as noted in the warning under maximum power dissipation, a high level of input overdrive in a high noninverting gain circuit can result in a large current flow in the input stage. though this current is internally limited to about 30 ma, its effect on the total power dissipation may be significant. high performance video line driver at a gain of +2, the ad8013 makes an excellent driver for a back terminated 75 w video line (figures 31, 32, and 33). low differential gain and phase errors and wide 0.1 db bandwidth can be realized. the low gain and group delay matching errors ensure excellent performance in rgb systems. figures 34 and 35 show the worst case matching. 75 w 75 w v out 75 w cable 75 w 75 w cable 4 +v s ad8013 0.1? 11 0.1? ? s r g v in r f figure 31. a video line driver operating at a gain of +2 (r f = r g from table i) frequency ?hz 1m 1g 10m closed-loop gain (normalized) ?db 100m ? +1 0 ? ? ? ? ? 0 ?0 ?80 ?70 phase shift ?degrees g = +2 r l = 150 w v s = 5v v s = +5v v s = +5v v s = 5v gain phase figure 32. closed-loop gain & phase vs. frequency for the line driver frequency ?hz 1m 1g 10m normalized gain ?db 100m +0.1 0 ?.1 ?.2 ?.3 ?.4 ?.5 g = +2 r l = 150 w v s = +5v v s = 5v +0.2 figure 33. fine-scale gain flatness vs. frequency, g = +2, r l = 150 w
C11C rev. a ad8013 frequency ?hz 1.5 1.0 ?.0 1m 1g 10m gain matching ?db 100m 0.5 0 ?.5 ?.0 ?.5 g = +2 r l = 150 w v s = +5v v s = 5v figure 34. closed-loop gain matching vs. frequency frequency ?hz 10 8 2 4 6 ?.0 0.5 0 ?.5 1.0 100k 100m 1m group delay ?ns 10m v s = +5v v s = 5v g = +2 r l = 150 w g = +2 r l = 150 w delay matching delay v s = +5v v s = 5v figure 35. group delay and group delay matching vs. frequency, g = +2, r l = 150 w disable mode operation pulling the voltage on any one of the disable pins about 1.6 v up from the negative supply will put the corresponding amplifier into a disabled, powered down, state. in this condition, the amplifiers quiescent current drops to about 0.3 ma, its output becomes a high impedance, and there is a high level of isolation from input to output. in the case of the gain of two line driver for example, the impedance at the output node will be about the same as for a 1.6 k w resistor (the feedback plus gain resistors) in parallel with a 12 pf capacitor and the input to output isolation will be about 66 db at 5 mhz. leaving the disable pin disconnected (floating) will leave the corresponding amplifier operational, in the enabled state. the input impedance of the disable pin is about 40 k w in parallel with a few picofarads. when driven to 0 v, with the negative supply at C5 v, about 100 m a flows into the disable pin. when the disable pins are driven by complementary output cmos logic, on a single 5 v supply, the disable and enable times are about 50 ns. when operated on dual supplies, level shifting will be required from standard logic outputs to the disable pins. figure 36 shows one possible method which results in a negligible increase in switching time. +5v 10k to disable pin v i v i high => amplifier enabled v i low => amplifier disabled ?v 4k 8k figure 36. level shifting to drive disable pins on dual supplies the ad8013s input stages include protection from the large differential input voltages that may be applied when disabled. internal clamps limit this voltage to about 3 v. the high input to output isolation will be maintained for voltages below this limit. 3:1 video multiplexer wiring the amplifier outputs together will form a 3:1 mux with excellent switching behavior. figure 37 shows a recommended configuration which results in C0.1 db bandwidth of 35 mhz and off channel isolation of 60 db at 10 mhz on 5 v supplies. the time to switch between channels is about 50 ns. switching time is virtually unaffected by signal level. 665 w 75 w v in 1 84 w 845 w disable 1 v out 75 w 75 w cable ? s 7 6 5 4 +v s 1 665 w 75 w v in 2 84 w 845 w disable 2 14 13 12 2 665 w 75 w v in 3 84 w 845 w 8 9 10 3 11 disable 3 figure 37. a fast switching 3:1 video mux (supply bypassing not shown) 10 0% 100 90 200ns 500mv 5v figure 38. channel switching characteristic for the 3:1 mux
ad8013 rev. a C12C c2084C18C10/95 printed in u.s.a. 2:1 video multiplexer configuring two amplifiers as unity gain followers and using the third to set the gain results in a high performance 2:1 mux (figures 39 and 40). this circuit takes advantage of the very low crosstalk between channels 2 and 3 to achieve the off channel isolation shown in figure 40. this circuit can achieve differential gain and phase of 0.03% and 0.07 respectively. v out v in a r1 2k w v in b r3 10 w r4 10 w r2 2k w r5 845 w r6 845 w 7 6 5 1 14 13 12 2 8 9 10 3 2 3 disable disable figure 39. 2:1 mux with high isolation and low differential gain and phase errors frequency ?hz 1g 1m closed-loop gain ?db 100m ? ? ? ? ? ? ? ? ?0 ?0 ?0 ?0 feedthrough ?db ?0 0 1 2 ?0 10m gain feedthrough figure 40. 2:1 mux on channel gain and mux off channel feedthrough vs. frequency gain switching the ad8013 can be used to build a circuit for switching between any two arbitrary gains while maintaining a constant input impedance. the example of figure 41 shows a circuit for switching between a noninverting gain of 1 and an inverting gain of 1. the total time for channel switching and output voltage settling is about 80 ns. 6 5 4 1 7 +5v dis 1 698 w 698 w 15 w v out 10 9 3 11 8 ?v dis 3 845 w 1k 845 w 1k 2k 13 14 12 50 w 100 w v in figure 41. circuit to switch between gains of C1 and +1 10 0% 100 90 200ns 500mv 5v 500mv figure 42. switching characteristic for circuit of figure 41 outline dimensions dimensions shown in inches and (mm). 14-lead plastic dip (n-14) 14 17 8 0.795 (20.19) 0.725 (18.42) 0.280 (7.11) 0.240 (6.10) pin 1 seating plane 0.022 (0.558) 0.014 (0.356) 0.060 (1.52) 0.015 (0.38) 0.210 (5.33) max 0.130 (3.30) min 0.070 (1.77) 0.045 (1.15) 0.100 (2.54) bsc 0.160 (4.06) 0.115 (2.93) 0.325 (8.25) 0.300 (7.62) 0.015 (0.381) 0.008 (0.204) 0.195 (4.95) 0.115 (2.93) 14-lead soic (r-14) 14 8 7 1 0.3444 (8.75) 0.3367 (8.55) 0.2440 (6.20) 0.2284 (5.80) 0.1574 (4.00) 0.1497 (3.80) pin 1 seating plane 0.0098 (0.25) 0.0040 (0.10) 0.0192 (0.49) 0.0138 (0.35) 0.0688 (1.75) 0.0532 (1.35) 0.0500 (1.27) bsc 0.0098 (0.25) 0.0075 (0.19) 0.0500 (1.27) 0.0160 (0.41) 8 0 0.0196 (0.50) 0.0099 (0.25) x 45

▲Up To Search▲

Price & Availability of AD8013ARZ-14

	To Download AD8013ARZ-14 Datasheet File
If you can't view the Datasheet, Please click here to try to view without PDF Reader .